Recent improvements for DC power supply circuits are increasingly important in portable and battery powered devices. In such devices, a supply voltage is used that is sometimes provided by an AC to DC transformer, or a “brick”, which outputs a DC voltage such as 12 or 24 Volts when AC power is available. Portable devices often also operate on similar DC voltages provided from rechargeable or other batteries when AC power is not available. Some portable devices may not have a “brick” but operate only from batteries. Electronics used within the portable devices typically include integrated circuits such as a microprocessor, volatile or non-volatile storage devices, digital radio or cellphone transceiver devices, and other functions such as Bluetooth, WiFi, and display drivers. The integrated circuit devices are increasingly designed to operate at lower and lower operating voltages, such as 1.8 Volts DC or even lower. Lower operating voltages for integrated circuits consume less power and thus extend battery life. Other supply voltages such as 2.8V, 3.3V, or 5 V are sometimes used. The system supply voltage from the batteries or the AC to DC transformer or “brick” is typically higher than the voltage needed by the electronic circuitry, so a DC-DC step down converter is used.
Switching power converter circuits are increasingly used to provide the DC voltage and current needed for electronic devices. In the case of a “step down” switching converter, pulse width modulated (“PWM”) converters in a “buck” configuration are often used. These PWM converter circuits are far more efficient and run cooler than the linear regulators used previously to provide the stepped down DC-DC voltage. In a buck converter, a high side switch (which may be, for example, a MOS transistor) is coupled with its current conduction path between an input voltage terminal and a switching node. A pulse width modulated signal coupled to a gate terminal of the high side switch is used to turn on or “close” the high side switch in an “on” state, and the pulse width modulated signal is used to turn off or “open” the high side switch in an “off” state. These two states alternate in a more or less constant frequency pattern. The “duty cycle” of the converter is a ratio of the “on” time of the high side switch to the “off” time. An inductor is coupled between the switching node and an output terminal for the output voltage. An output capacitor is coupled between the output terminal and a ground terminal. By closing the high side switch for the “on” state time, and driving current into the inductor during the “on” state, and then subsequently opening the high side switch for the “off” state time, current flows into the inductor and into the load, and an output voltage is developed across the load that is supported by the output capacitor. A rectifying device is also provided coupled between the switching node and a ground potential. The rectifying device is used to supply current into the inductor when the high side switch is open, the “off time” for the circuit. Increasingly this rectifying device is replaced by a low side driver switch; although diode rectifiers are sometimes used. Use of a MOSFET transistor for both the high side switch and the low side switch (replacing the older diode rectifier) creates a synchronous switching converter topology. By using MOSFET transistors with low RDSon values, and by controlling the on and off times for the high side and low side switches, efficient DC-DC buck converter circuits are implemented.
As is known to those skilled in the art, in a switching buck converter that uses a constant frequency and a duty cycle with pulse width modulation, when output current is constantly flowing to the load, the DC output voltage obtained at the output terminal is directly proportional to the input DC voltage at the voltage input terminal. More specifically, the output voltage is proportional to the input voltage multiplied by the ratio of the high side switch on-time to the off-time, that is, the DC output voltage is proportional to the duty cycle. Thus by changing the pulse width of the “on” state, the output voltage may be varied to a desired value, and it may be regulated. An onboard or off board oscillator is typically used to obtain a pulse source which clocks the circuit. Using sense resistors or other current sensors at the output along with feedback control, for example, the output voltage can then be regulated to a desired value by varying the width of the modulated pulse that closes the high side switch, thereby coupling the input voltage or a supply voltage to the switching node for the inductor. Additional circuitry is sometimes used to regulate the output during times when no current or low current is flowing into the load, for example the circuit may switch to a pulsed frequency mode or otherwise skip cycles when light load conditions are present. As an example, U.S. Pat. No. 8,710,816, issued Apr. 29, 2014, entitled “Buck Converter having Reduced Ripple under Light Load”, to Miyazaki, which is co-owned with the present application, and which is hereby incorporated herein in its entirety by reference, discloses circuitry for increasing the efficiency of a buck converter circuit when operating under light load.
While the buck converter is substantially more efficient than the linear regulators used in the past to provide DC voltages, multiple phase buck converters are increasingly used to still further improve buck converter performance. In a multiple phase buck converter, several switching circuit stages and corresponding inductors are coupled in parallel to one another, and these multiple stages are operated in non-overlapping phases. The multiple phase outputs are then simply added to form the overall output. There may be two, three, four or more phases and corresponding circuit stages. However the addition of the multiple phases increases the complexity of the control circuitry, and a design trade-off thus exists between the number of phases used and the amount of and complexity of the control circuitry required.
Use of multiple phase converters advantageously decreases the undesirable ripple voltage at the output observed for single phase buck converters, and a multiphase buck converter also handles variations in the load current very well when compared to a single phase buck converter. Since modern microprocessors have many “sleep” and “power save” modes that are used to reduce the power used during idle microprocessor cycles and so extend the battery life of portable devices, the current demanded by a modern microprocessor will vary substantially. Multiphase buck converters are therefore increasingly used, particularly for supplying DC voltages in microprocessor systems.
FIG. 1 depicts, for the purpose of explanation, a block diagram of a typical multiphase buck converter circuit 10. In FIG. 1, a first stage switching circuit 11 is shown with a high side MOSFET switch 111, which is an N-type MOSFET that is sufficiently large to provide the required or expected load current to the corresponding inductor L_1 during the “on” state. A high side driver circuit 113 is provided coupled to the gate terminal of the MOSFET switch 111. The high side MOSFET switch 111 is coupled to switching node SW1 which is coupled to one terminal of inductor L_1. Further, in first stage switching circuit 11, a low side switch 117, which in this example is also an N type MOSFET device 117, is coupled between the switch node SW1 and a ground terminal. Low side driver 115 controls low side switch 117 by controlling the voltage on the gate terminal of the low side switch 117. During the “off” state of the switching circuit 11, the low side switch 117 provides a current path to supply current to the inductor L_1.
In FIG. 1 the multiphase buck converter 10 has n phases, as indicated by the asterisks. In the example depicted, two phases are shown. However, in practical systems n can be any positive integer greater than or equal to two, and three and four and more phase buck converter systems are known for various applications. This is indicated in FIG. 1 by the asterisks in the column between the first stage inductor L_1 and the inductor for the bottom stage, labeled L_N.
In FIG. 1, the second stage switching circuit 13 is shown coupled in parallel to the first stage switching circuit 11. The circuit elements within second stage switching circuit 13 are duplicated from first stage switching circuit 11 and include a high side MOS switch 131, which again can be an N-type MOSFET transistor, a high side driver circuit 133 coupled to the gate terminal of the high side MOS switch 131, and a low side driver circuit 135 which is coupled to the gate terminal of the low side switch 137. The switching circuit 13 is coupled to a switching node SWN which is coupled to one terminal of the inductor L_N.
Driver control circuit 15 in FIG. 1 provides the control of the high side driver circuits 113 and 133, and of low side driver circuits 115 and 135. In operation, in a first phase the high side MOS switch 111 is closed by driving a gate voltage onto the gate terminal from the high side driver circuit 113 that exceeds the source voltage by a transistor threshold voltage Vt for the high side MOS switch, transistor 111. This action “closes” the high side MOS switch 111 and couples the input voltage Vin to the switching node SW1. Current flows into the inductor L_1 and out to the output node, charging capacitor CO and the load current flows forming an output voltage at the output Vo. During this “on” state, inductor L_1 stores energy in a magnetic field surrounding the inductor. After the “on” state ends, the driver control circuit 15 controls the high side driver circuit 113 and turns off the high side MOSFET switch 111, and the driver control circuit 15 controls low side driver circuit 115 and turns on the low side switch, MOSFET 117. The low side switch 117 provides a current path during the “off” state of the first stage switching circuit 11, so that current flows through the inductor L_1 from the stored energy and into the capacitor CO and into the load (not shown) at the output terminal Vo, thus supporting the voltage at the output terminal Vo during the “off” state.
The second stage switching circuit 13 is operated in the same fashion as the first stage switching circuit 11, but the two stages are operated in non-overlapping phases. In this fashion, the output current provided by the two phase switching circuits 11 and 13 is added at the output Vo and together the two switching circuits 11 and 13 provide the current to the load. Driver control circuit 15 provides the pulses needed to turn on the high side driver circuits 113 and 133 for the high side MOS switches 111 and 131, and to turn on the low side switches 117 and 137, in non-overlapping phases.
In order to turn on the high side MOS switches 111 and 131, a voltage at the gate terminal of the MOS transistors that is higher than the input voltage is needed. In prior known approaches, this gate voltage has been formed using a bootstrap capacitor. Sometimes this capacitor is referred to as a “fly cap”, but for the descriptive purposes of this disclosure, the term “bootstrap capacitor” is used. The bootstrap capacitor is first configured with a top plate coupled to a positive supply voltage, such as an internally regulated voltage Vdd, and a bottom plate coupled to a ground potential. In this manner the bootstrap capacitor is charged to the supply voltage level. The bootstrap capacitor is later coupled so that the bottom plate is at the positive input voltage VIN and the top plate is coupled to the high side switch gate. The voltage at the high side switch gate is thus “bootstrapped” to a voltage that is the sum of the positive supply voltage and the input voltage VIN at the bottom plate. In this manner the gate voltage needed for turning on the high side MOS switches 111, 131 can be developed.
In a conventional multiple phase buck converter using a bootstrap capacitor to provide the needed gate voltage at the high side MOS switch, each switching circuit stage needs an individual bootstrap capacitor. Further, the high side switch devices, which are N-type MOSFET transistors, have a large gate capacitance. The bootstrap capacitor required for each stage is therefore also quite large as it needs to charge the gate capacitor of the high side MOS switch. The use of a multiple phase buck converter configuration as known in the prior art thus requires multiple large bootstrap capacitors. If these bootstrap capacitors are integrated with the high side switches and low side switches in a converter integrated circuit, the amount of silicon area required for the bootstrap capacitors may make the multiple phase buck converter circuit too large to be fabricated on a single device that is to be manufactured in a certain semiconductor process. Alternatively, if the bootstrap capacitors are instead provided as external components coupled to an integrated circuit, the use of the multiple bootstrap capacitors requires two external pins for each of these added components. The extra pins which may undesirably increase pin count for the converter integrated circuit, and correspondingly, increase packaging and other manufacturing costs. This may lead to a situation where the needed pins are simply not available. Further the use of multiple large external bootstrap capacitors undesirably increases the board area required for implementing the multiple phase DC-DC buck converter.
Improvements in multiple phase buck converters are therefore needed to address the deficiencies and disadvantages of the known prior approaches.